On the Electrodynamics of a Non-Contact Inductive Sensor for High-Accuracy Tip Deflection Measurements of Piezoceramic Bending Actuators and its Application for Energy Harvesting Purposes

Introduction

The piezoelectric effect combines electrical and mechanical parameters. The conversion of a mechanical deformation into an electrical signal is called direct piezoelectric effect, the resulting deformation by applying an electric field is referred to as the reciprocal piezoelectric effect. These effects can be used for piezoceramic sensors and actuators offering a wide range of applications in the fields of acoustics, automation, communication, health care, automotive and consumer products.

The development of low drift piezoceramics reduces the effects of relaxation and hysteresis, but their low piezoelectric coefficients do not allow their use as high-efficient piezoceramic actuators. So, the range of applications with low drift piezoceramics is limited due to small deflections and forces. For larger deflections and forces a smart sensor-actuator-system is required. This paper introduces a high efficient piezoceramic bending actuator and an attached non-contact inductive proximity sensor which can be implemented into a closed control loop. The usage of the attached sensor in combination with a high-accuracy electronic circuit allows the detection and compensation of disturbing influences like inherent piezoelectric effects, external forces, moments and vibrations [1].

Inductive Sensor Principle

Figure 1 shows the measurement setup of the non-contact inductive proximity sensor for tip deflection measurements of piezoelectric bending actuators.

The non-contact inductive proximity sensor consists of an application specific integrated circuit (ASIC) and two external 3D-microcoils, which are connected to the ASIC. One coil acts as sensing coil, the other one provides a reference. The sensing coil is connected to the integrated oscillator 1. Its resonant frequency $f_1$ changes during the approach of the actuator’s tip which is covered by a detectable Cu-layer target (see Figure 2).

If the sensing coil is driven by an alternating current $i_1$, an alternating magnetic field will be generated. In its turn, this causes eddy currents within the Cu-layer generating magnetic fields, that counteract the external magnetic field (LENZ’s law). As a consequence, the external magnetic field decreases, resulting in a decrease $\Delta L_1$ of the inductance $L_1$ of the sensing coil. Thus, the changing resonant frequency of the oscillator 1 is a measure for the deflection $\zeta$ of the bender’s tip. The reference coil is connected to the integrated oscillator 2. Its arrangement prevents an influence by the Cu-layer. With its constant inductance $L_2$, the reference coil provides the generation of the reference frequency $f_2$. The difference $|f_1-f_2|$ and the sum $|f_1+f_2|$ of both frequency signals are evaluated by means of the mixer and is fed through the following lowpass filter. The lowpass filter suppresses deterministically caused higher-frequency signals. The following Schmitt trigger provides a shaping of both difference frequency signals, thus the output signal $f_{\mathrm{out}}$ is available in binary form (TTL level) [2].

Both oscillator circuits consist of an integrated capacitance $C$ as well as an active cell, maintaining the oscillation. The sensing and reference coil complete the respective $L$-$C$-resonant circuit. In both $L$-$C$-resonant circuits, the capacitance $C$ can be assumed to be equal. The frequency $f_2$ of the resonant circuit 2 is specified by the reference coil $L_2$ according to

If the conductive Cu-layer target approaches the sensing coil, eddy currents are generated within the layer weakening the alternating magnetic field of the sensing coil. Thus, the effective inductance $L_{\mathrm{eff}}=L_1-\Delta L_1$ of the sensing coil decreases and the oscillator’s resonant frequency increases from $f_1$ to

The general correlation between the relative frequency change and the relative inductance change concerning the oscillator 1 yields in combination with $f_{\mathrm{eff}}=f_1+\Delta f_1$

Since the relative inductance change $\Delta L_1$ amounts up to $-30\%$ during approximation of the Cu-layer target from a distance of several centimeters to almost contact with the coil, the maximum relative frequency change amounts 15% with respect to Eq. (3). The mixer evaluates the difference of both frequencies $f_{\mathrm{eff}}$ and $f_2$. In order to achieve a reliable and a non-disturbed signal on TTL level, the difference signal is filtered by a lowpass filter and shaped by a Schmitt trigger. Thus, the output frequency results in

Dependent on $\Delta L_1/L_1$ of the oscillator 1, the relative change of the output frequency yields in combination with Eq. (4):

Equation (5) shows a very high relative output frequency change being achieved, if both nominal inductances of the sensing and reference coil are nearly equal, even though the change of $L_1$ and thus, the distance change between sensing coil and Cu-layer target can be very small. This aspect constitutes the choice of an inductive proximity sensor for high-accuracy tip deflection measurements for a piezoelectric bending actuator.

Equivalent Network Representation

The equivalent network of the sensing coil close to the conductive Cu-layer is illustrated in Figure 3a [3]. It is assumed, that the Cu-layer carrying the eddy current $i_s$, can be represented as a series connection of the inductance $L_s$ and the resistance $R_s$. The sensing coil is represented by the inductance $L_1$ and the resistance $R_1$. Due to the coupling between both $L$-$R$-circuits, an alternate circuit representation is to be achieved, which allows for the representation of the arising effective inductance $L_{\mathrm{eff}}$ as a function of the distance $z$ between sensing coil and Cu-layer (see Figure 3b).

In order to describe the interaction between the Cu-layer and sensing coil, an equivalent circuit representation of a transformer without iron core is considered (see Figure 4). Assuming two coils with $N_1$ and $N_2$ turns and taking LENZ’s law into account, following pair of equations can be formulated for both meshes of the circuit shown in Figure 4:

The quantities $\Phi_1$ and $\Phi_2$ correspond to the magnetic flux correlated with coil 1 and 2, respectively. The individual fluxes are proportional to the currents $i_1$ and $i_2$:

In the Eqs. (8) and (9) the quantities $L_1$ and $L_2$ denote the self-inductances, the quantity $M$ denotes the mutual inductance. Taking Eqs. (8) and (9) into consideration, both Eqs. (6) and (7) can be reformulated:

The correlations formulated in Eqs. (10) and (11) are applied to the equivalent circuit representation in Figure 3a. Following analogies are achieved:

Using the analogies in Eq. (12), the Eqs. (10) and (11) can be reformulated according to

In case of steady state currents and voltages, the complex root mean square values yield:

Rearranging of Eq. (16) with respect to the eddy current $I_s$ and following insertion into Eq. (15) results in the potential difference at the clamps of oscillator 1:

Due to the high conductivity of the Cu-layer target attached to the bender’s tip ($R_s^2 \ll\omega^2 L_s^2$), Eq. (17) can be simplified:

Thus, the resistance $R_e$ and the inductance $L_e$ of the equivalent network in Figure 3b are known. The definition of the coupling coefficient

implies the formulation of the mutual inductance $M(z)$ as a function of distance $z$ between the bender’s tip and sensing coil. Thus, the equivalent quantities $R_e$ and $L_e$ are dependent on the distance $z$ according to

and

Equation (21) allows the formulation of the relative inductance change of the sensing coil 1:

With respect to Eq. (1), the basic inductance $L_1(z \to\infty)$ is correlated with the frequency $f_1$ of oscillator 1:

Similarly, the equivalent inductance yields

Thus, Eq. (22) can be reformulated:

In a next step, the distance-dependent relative inductance change will be calculated by an electrodynamic approach.

Inductance of a Circular Loop Influenced by a Conductive Layer

The arrangement of a circular loop opposite to a conductive layer is shown in Figure 5. With respect to the origin of the coordinate system, the circular loop is parallel to the surface layer in distance $z=-d$. The represented arrangement is axially symmetric, thus it is described by means of cylindrical coordinates $(r,\theta,z)$.

With respect to MAXWELL’s equations for time-variable fields, the electric field E generates a magnetic field H by means of the conduction and displacement current. On its part, the magnetic field H generates an electric field by means of induction. Thus, electric and magnetic fields are connected with each other. For the physical description of the arrangement illustrated in Figure 5, the vector of the magnetic field strength $\mathbf{H}(r,t)$ is of interest.

Solution Approach

The fundamental describing MAXWELL equations are:

Taking the material-specific relations for the electric displacement $\mathbf{D}(r,t)$, the current density $\mathbf{J}(r,t)$ and the magnetic induction $\mathbf{B}(r,t)$ in accordance with

into account, it can be written:

The permittivity $\varepsilon$, the conductivity $\kappa$ and the permeability $\mu$ in Eqs. (31) and (32) are assumed to exhibit a time-independent, isotropic behavior. The application of the vector operation "curl" to Eq. (32) and following insertion of Eq. (31) yields:

According to the calculation rules of vector analysis, it can be written in general [4]:

Thus, Eq. (33) can be reformulated:

Since there is no source of magnetic field ($\operatorname{div}\mathbf{H}(\mathbf{r},t)=0$), Eq. (35) can be simplified:

Equation (36) results in two differential equations, one for the metallic semi-infinite space ($\varepsilon=0$, $\mu$, $\kappa$) and one for the medium air ($\varepsilon=\varepsilon_{0}$, $\mu=\mu_{0}$, $\kappa=0$):

The expressions (37) and (38) represent the electromagnetic skin and wave equation concerning the magnetic field in metal and air, respectively. Since cylindrical coordinates are used, the Laplacian $\Delta$ has to be formulated explicitly. Taking the angle independence of the magnetic field ($\partial\mathbf{H}(\mathbf{r},t)/\partial\theta=0$) into consideration, the necessary differential equations of the $z$-component of the magnetic field result in their final formulation [7]:

Alternating Magnetic Field Within a Conductive Layer

In order to find a solution for the alternating magnetic field within the conductive layer, Eq. (39) is reformulated according to:

The separation of variables serves as method for finding a general solution:

Since the circular loop is subjected to a sinusoidal alternating current $I(t)=I e^{j\omega t}$, the real time function $H_z(r,z,t)$ is replaced by the complex time function $H_z(r,z,t)$. Thus, Eq. (42) can be written as follows:

Insertion of Eq. (43) into Eq. (41) and following separation of the variables $(r,z,t)$ yields

Here, $\nu$ is a constant, since the left side of Eq. (44) is independent of $t$ and the right side is independent of $r$ and $z$. Applying the method of separation of variables again results in

Insertion of Eq. (45) into Eq. (44) yields

where $u_n$ denotes another constant. Two ordinary differential equations for $X(r)$ and $Y(z)$ result from Eq. (46):

The solution $Y(z)$ of the homogeneous linear differential equation Eq. (47) results from the linear combination of two linear independent solutions (basis solutions). Here, the solutions $e^{\lambda z}$ and $e^{-\lambda z}$ represent the fundamental system of the differential equation. Following solution is achieved:

In order to find a solution $X(r)$ for the differential equation Eq. (48), a transition to the variable $\rho:=u_n r$ is defined. With the definition

and following insertion into Eq. (48), BESSEL’s differential equation follows:

The general solution of Eq. (51) results in [5]:

Choosing $C_1 = 1$, which does not influence the total solution at all, and taking Eq. (50) into consideration, the solution $X(r)$ is achieved:

The expression $J_0(u_n r)$ represents BESSEL's function of first kind and order zero. Taking Eqs. (43) and (41) into account, the formal solution is achieved by superposition [5]:

Due to the continuous spectrum of the eigenvalues $u$, Eq. (54) can be written in integral form [6]:

Finally, the quantities $A_1$ and $A_2$ can be solved by means of corresponding boundary conditions.

The general solution for the alternating magnetic field in the air is similar to Eq. (55), while the term $j\omega\kappa\mu$ has to be replaced by the term $-\varepsilon_0\mu_0\omega^2$. Since the oscillator 1 of the available inductive proximity sensor exhibits an oscillation frequency of $f \approx17\,\mathrm{MHz}$, the term $-\varepsilon_0\mu_0\omega^2$ can be neglected. Thus, the alternating magnetic field in the air is formulated as following:

Magnetic Field of a Circular Loop

The geometrical quantities being necessary for the calculation of the magnetic field of a circular loop (radius $a$) along its axis in distance $z$ from the center (point of origin) are shown in Figure 6. The vector $\mathrm{d}l$ points in direction of the alternating current $I(t)=Ie^{j\omega t}$. The current loop in space is characterized by the quantity $C$, the vector $\mathbf{r}_d$ points from the source point Q to the space point P. Due to the present symmetry, cylindrical coordinates are used for following considerations.

In order to calculate the magnetic field of a circular loop along its axis of symmetry, the BIOT-SAVART law is considered:

The vector $\mathrm{d}\mathbf{l}$ can be defined using the unit vector $\mathbf{e}_{\varphi}$:

The vector $\mathbf{r}_{\mathrm{d}}$ pointing from source point Q at space point P results from the difference of the position vectors $\mathbf{r}_{\mathrm{p}}$ and $\mathbf{r}_{\mathrm{q}}$:

The vector product $\mathrm{d}\mathbf{l}\times\mathbf{r}_{\mathrm{d}}$ can be determined with Eqs. (58) and (59).

Furthermore, Eq. (59) yields

Insertion of Eqs. (60) and (61) into Eq. (57) yields the axial component of the magnetic field:

In order to provide a general representation of the magnetic field of a circular loop in the following form

following integral has to be evaluated [8]:

BESSEL’s function of first kind and order zero can be defined by a complex exponential function (generating function) in integral form [9, 10]:

Insertion of Eq. (65) into Eq. (64) results in

Rewriting (*) in the form

inserting it into Eq. (66) and integrating it in combination with appropriate integral tables [11] gives following correlation

The derivative with respect to the variable $a$ according to

together with the general derivation rule for BESSEL’s function [12]

results in the expression

Multiplication with $Ie^{j\omega t}/2$ on both sides of Eq. (71) finally results in

which exactly matches the analytical description (62) of the magnetic field of a circular loop along the $z$-axis.

Since the origin of the coordinate system in Figure 5 is not located in the center of the circular loop, but shifted in $z$-direction ($z=-d$), Eq. (72) results in

The expression defined in Eq. (73) has to be equivalent to the general formulation of the magnetic field in Eq. (63) with respect to $r=0$, thus with $J_0(0)=1$ following condition can be formulated:

Comparing left and right side of Eq. (74) immediately yields

resulting in

Comparing left and right side of Eq. (76), the constant $B_1$ can be formulated according to

Inserting Eq. (77) into Eq. (63) and taking Eq. (75) into consideration, the general description of the magnetic field of a circular loop dependent on the coordinates $(r,z,t)$ is achieved:

Equation (78) allows for calculation of the magnetic field in the plane of the circular loop ($z=-d$).

The achieved expression in Eq. (79) will play an important role for the analytical calculation of the relative inductance change of a circular loop in presence of a conductive layer.

Influence of a Conductive Layer

In order to investigate the influence of a conductive layer of thickness $h$ on a circular loop, the upper and lower surface of the conductive layer are assumed to be formed by the two planes at $z=0$ and $z=h$ (see Figure 5). For the consideration of the influence on the space $z<0$, the magnetic field of the circular loop Eq. (78) has to be modified by an additional term. This will ensure the analytical formulation corresponding to the general form in Eq. (63) and moreover, the part of the additional magnetic field is zero for $z\to-\infty$. The magnetic field $H'_z$ between circular loop and conductive layer ($-d<z<0$) can be described by

where $B_1$ has already been defined by Eq. (77). $B_2$ has to be determined.

Within the conductive layer, the description of the magnetic field results from Eq. (55). Using the definition

for ($0<z<h$), expression Eq. (55) is reformulated according to

The magnetic field outside the layer ($z>h$) can be described similarly to the expression Eq. (63). However, the positive exponential term has to be excluded:

In order to determine the four unknown quantities $B_1$, $A_1$, $A_2$ and $C_1$, the continuity characteristics of the normal component $B_n$ of the magnetic induction and the discontinuity characteristics of the normal component $H_n$ of the magnetic field are consulted with respect to both boundary surfaces of the conductive layer at $z=0$, $z=h$ and $r=0$, i.e.

and

The following system of equations is achieved:

Since the magnetic field between circular loop and conductive layer is of interest, the quantity $B_2$ has to be determined. Eliminating of $A_1$, $A_2$ and $C_1$ yields:

The axial component of the magnetic field within the plane of the circular loop $H'_z(r,z=-d,t)=H'_{\mathrm{coil}}$ in presence of the conductive layer yields with Eq. (80)

where $B_1$ and $B_2$ are defined by Eq. (77) and Eq. (90), respectively.

Relative Inductance Change of a Circular Loop in Presence of a Conductive Layer

On the basis of the definition for the magnetic flux within the plane of the circular loop in accordance with

a conductive layer influences the magnetic field of the coil $H_{\mathrm{coil}}$ resulting in a change of the magnetic flux:

Insertion of Eqs. (79) and (91) into Eq. (93) yields the change of the magnetic flux at position $z=-d$

and thus

Insertion of definition (77) for $B_1$ into Eq. (95) results in

The evaluation of the inner integral results in [13]:

Thus, Eq. (96) can be written in following form:

Due to the change of the magnetic flux, a voltage $\Delta U$ is induced resulting in

Insertion of Eq. (98) into Eq. (99) yields the impedance change $\Delta Z$:

The inductance of an idealized, free circular loop with small wire radius $r_{\mathrm{w}}$ compared to the loop radius $a$ can be represented according to [7]:

Thus, from Eq. (100) follows

Introducing the dimensionless variable

(102) can be written as

Comparing the left side of Eq. (104) with the right side gives evidence, that the quotient of both integration constants $B_{1}$ and $B_{2}$ has to be formulated as a complex number

In order to calculate the real part $\alpha(z^{*})$ and the imaginary part $\beta(z^{*})$, a representation of the complex root $\sqrt{1+jk}$ in Eq. (90) is needed as a complex number by means of the following relationship [8]:

Since the representation of the quotient in Eq. (90) as a complex number according to (105) comes along with an enormous mathematical calculation complexity, the software package Mathematica was used, in order to ease calculation significantly. Furthermore, the relative inductance change in Eq. (104) is of interest, thus a representation of $\beta(z^{*})$ in Eq. (105) is set aside.

In combination with Eq. (90), Eq. (105) and Eq. (106), the fundamentals for the calculation of the relative inductance change of a circular loop in presence of a conductive layer are finally achieved:

where

with

and

It should be noted, that Eq. (107) is not solvable in closed form analysis, thus a numeric integration has to be considered.

Measurement Setups and Results

Relative Inductance Change vs. Tip Deflection

The relative inductance change of the sensing coil in presence of a conductive Cu-layer applied to the bender’s tip is experimentally determined as a function of the distance $z$. The measurement results are compared with the results of the analytical calculations based on the fundamental Eq. (107).

The determination of the relative inductance change of the sensing coil with respect to the conductive Cu-layer is realized with the measurement setup shown in Figure 7.

Since the geometry of the arrangement plays a fundamental role concerning the analytical calculations, the circle sector sketched in Figure 7 is represented in higher magnification in Figure 8 showing the non-deflection and deflection state of the piezoelectric bending actuator.

Here, $z_0$ denotes the maximum distance between the upper side of the Cu-layer and the central position of the sensing coil for the non-deflection state of the bending actuator, $z$ denotes the distance between the upper side of the Cu-layer and the central position of the sensing coil for the deflection state of the bending actuator, $\tilde{\xi}_{\max}$ corresponds to the maximum deflection of the bending actuator, $b$ corresponds to the distance between the front edges of the coil and the ceramic carrier, $a$ denotes the coil radius, $l$ corresponds to the coil length and $d_{\mathrm{air}}$ denotes the air gap between ceramic carrier and Cu-layer at maximum deflection. The physical dimensions of the quantities $\tilde{\xi}_{\max}$, $b$, $a$, $l$ and $d_{\mathrm{air}}$ being necessary for the analytical calculations of the relative inductance change of the sensing coil in presence of the Cu-layer are listed in Table 1.

$\tilde{\xi}_{\max}$ / $\mu\mathrm{m}$	$b$ / $\mu\mathrm{m}$	$a$ / $\mu\mathrm{m}$	$l$ / $\mu\mathrm{m}$	$d_{\mathrm{air}}$ / $\mu\mathrm{m}$
329.6	341.2	513.0	600.0	15.0

Taking Figure 8 into consideration, the following correlation can be formulated for the deflection state of the bending actuator:

Furthermore, for an arbitrary deflection $0 < \tilde{\xi} < \tilde{\xi}_{\max}$, it can be written

thus, the distance $z$ between Cu-layer and central position of the sensing coil can be calculated by insertion of Eq. (113) into Eq. (112):

Taking the Eqs. (107)–(111) into consideration, the relative inductance change is analytically calculated. The variable $d$ in the Eqs. (107)–(109) and (111) has to be replaced by the variable $z$ in accordance with Eq. (114).

The starting point for the experimental characterization of the relative inductance change is represented by following equation defined already in Eq. (25):

Equation (115) shows, that both, the knowledge of frequency $f_1$ of the uninfluenced coil ($z \to\infty$) and frequency $f_e(z)$ of the coil affected by the Cu-layer in distance $z$ is necessary. The frequency measurements concerning $f_1$ and $f_1(z)$ are achieved with an oscilloscope. Since the distance $z$ is directly correlated with the deflection $\xi$ of the bending actuator with respect to Eq. (114), the relative inductance change of the sensing coil can be represented as a function of the deflection $\xi$ of the bending actuator. Figure 9 shows the experimentally determined and analytically calculated relative inductance change of the sensor-actuator-system as a function of the tip deflection $\xi$.

Tip Deflection vs. Applied Driving Voltage

Since it is also an aim to detect the hysteresis of the piezoelectric bending actuator by means of the integrated proximity sensor, the tip deflection has been measured in accordance with the measurement setup shown in Figure 10.

The amplifier stage following the function generator applies a sinusoidal voltage $V_{\mathrm{piezo}}$ with the frequency $f = 200\,\mathrm{mHz}$ to the piezoceramic bending actuator resulting in a bending movement of the cantilever beam. Thus, the distance between the copper layer and the sensing coil decreases and increases and the deflection $\xi$ of the bending actuator is transformed to a frequency signal $f_{\mathrm{out}}$ by the ASIC. In a next step, the frequency signal is converted to a voltage signal $V_{f/V}$ by a frequency-to-voltage converter and afterwards read in by an I/O board. The deflection measurement always verified by a high precision laser deflection measurement system provides a comparison and bench mark for the inductive proximity sensor in combination with the frequency-to-voltage converter [14]. Figure 11 shows the tip deflection $\xi$ versus the electric field $E$ (and thus the applied voltage $V_{\mathrm{piezo}}$). The measurement results show the ability of the inductive proximity sensor in combination with the appropriate sensor electronics to detect the hysteresis being typical for piezoceramic bending actuators.

Deflection Measurement for Energy Harvesting Purposes

Measurement of the deflection $\xi$ of a cantilever bending actuator allows to derive the amount of power dissipation for energy harvesting applications. Piezoceramic bending actuators can be used as kinetic energy harvesters and are modeled as second-order spring mass damper system in a simplified approach, see Figure 12.

Further simplification is necessary to derive the governing equation for displacement of the amplitude of the system: it is assumed, that the mass of the excitation source is much larger than seismic mass $m$ and that the excitation source is practically not damped by the energy harvester. Thus, the excitation source is assumed to provide infinite power. An external vibration which is assumed to exert a sinusoidal force on the harvester frame, causes the frame to move harmonically with

where $\hat{y}$ indicates the amplitude of the frame motion, and $\omega$ denotes the angular vibration frequency. The relative movement of the seismic mass $m$ with respect to the frame is given by

where $\xi$ indicates the amplitude of the mass motion and thus the bender’s tip deflection to be measured with the inductive proximity sensor. Since both, the frame motion and the motion of mass represent harmonic quantities, the Eqs. (116) and (117) can be expressed by complex amplitudes according to

and

Using complex amplitudes, the model of the kinetic energy harvester shown in Figure 12 can be transformed into a mechanical scheme, see Figure 13a. Due to electromechanical feedback, the transducer exerts a restoring force $F_{\mathrm{e}}$ on the seismic mass if an interface circuit is connected, causing additional electrical damping $d_{\mathrm{e}}$. The sum of the mechanical damping $d$ and the electrical damping $d_{\mathrm{e}}$ is defined by the quantity $\tilde{d}$, see Figure 13b.

In a next step, the mechanical scheme shown in Figure 13b is transferred to the equivalent circuit representation according to Figure 14. Here a mechanical force is represented as an electric current, and a mechanical velocity can be replaced as an electrical voltage. The mass is represented as a capacitor with the value $m$, the spring is modelled as an inductor with the value $c$ (spring compliance) and the sum of the mechanical and the electrical dampings is described by the resistor $\tilde{d}$. The voltage source models the external vibration.

Taking the mechanical admittances $h_1$ and $h_2$ in Figure 14 into account, the ratio between the bender’s tip deflection $\underline{\xi}$ and the movement $\underline{y}$ of the frame results in:

with the resonance frequency

and the quality factor

The displacement amplitude of the seismic mass $m$ can be written as:

Thus, the amplitude $\hat{y}$ of the frame motion can be determined by measuring the bender’s tip deflection $\hat{\xi}$.

In order to determine the power dissipated in the transducer, i.e. in the resistor $d_e$, Figure 15 is considered.

It can be written:

The velocity $v_1$ is gained by the the velocities of the frame an the bender’s tip and thus by the amplitudes of the frame motion and the tip deflection:

Since the amplitude of the frame motion can be determined by means of measuring the tip deflection according to Eq. (123), also the quantity $v_1$ and thus the power dissipated in the transducer according to Eq. (124) can be determined.

Conclusion

In this paper, we focused on the electrodynamics of a non-contact inductive proximity sensor for high-accuracy tip deflection measurements of piezoelectric bending actuators. The sensor principle itself is based on eddy current effects, offering the possibility of sensor integration in smallest spaces in combination with a high-accuracy electronic circuit. In order to determine the relative change of the inductance as a function of the beam bender’s tip deflection in an analytical way, an electrodynamic approach was formulated verifying the experimental measurement results. Besides the ability to detect the hysteresis, which represents a typical characteristic of piezoceramic bending actuators, the measurement of the displacement via a non-contact inductive sensor system has also the potential to directly evaluate the possible amount of energy which can be harvested from a piezoelectric energy harvester. In future, optimization of material and mechanical shape may be investigated and instantly evaluated in means of performance by measurement of the deflection and the external equivalent electrical parameters.

References